Method and apparatus for performing chip level equalization using joint processing

ABSTRACT

A method and apparatus for performing chip level equalization (CLE) using joint processing to enhance performance and system throughput using a transmitter having a plurality of transmit antennas and a receiver having a plurality of receive antennas. A channel response matrix is formed between the transmit antennas and the receive antennas to generate a joint channel correlation matrix between the transmit antennas and the receive antennas using a block-FFT (B-FFT) decomposition of the channel response matrix. Estimates of transmitted chip sequences from each of the transmit antennas are generated using minimum mean square error (MMSE) and the joint channel correlation matrix are combined. The combined estimate of the transmitted chip sequences are despread to recover transmitted data.

CROSS REFERENCE TO RELATED APPLICATIONS

This application claims the benefit of U.S. Provisional Patent Application Nos. 60/636,345 filed Dec. 14, 2004 and 60/642,383 filed Jan. 7, 2005, which are incorporated by reference as if fully set forth.

FIELD OF INVENTION

The present invention is related to a wireless communication receiver. More particularly, the present invention relates to a receiver that processes space-time transmit diversity (STTD), closed loop transmit diversity for transmit adaptive antennas and receiver diversity with over-sampling and fast Fourier transform (FFT)-based chip level equalization (CLE) using joint processing.

BACKGROUND

CLE is a candidate for use in advanced receivers in wireless communication systems for high data rate services such as high speed downlink packet access (HSDPA). CLE-based receivers, such as those used in wireless transmit/receive units (WTRUs), are used more often than Rake receivers in advanced receivers due to their superior performance.

Receive diversity using two or more receive antennas provides high performance by improving the reception quality of signals. Over-sampling is also used to improve the reception performance by rectifying performance degradations caused by timing errors or sampling errors. In addition, transmit adaptive antennas are used to improve signal degradations caused by fading, and thus improve data detection performance at the receiver and enhance the system throughput.

In conventional receivers which implement equalization, each channel that corresponds to an antenna is equalized independently of other channels that correspond to other antennas. However, these type of receivers usually experience significant performance degradations due to mutual channel interference from one antenna to another that cannot be eliminated or cancelled. Therefore, there is a need for receivers which implement CLE such that mutual channel interference is reduced or eliminated.

SUMMARY

The present invention is related to a method and apparatus for performing CLE using joint processing to enhance performance and system throughput using a transmitter having a plurality of transmit antennas and a receiver having a plurality of receive antennas. A channel response matrix is formed between the transmit antennas and the receive antennas to generate a joint channel correlation matrix between the transmit and the receive antennas using a block-FFT (B-FFT) decomposition of the channel response matrix. Estimates of transmitted chip sequences from each of the transmit antennas are generated using minimum mean square error (MMSE) and the joint channel correlation matrix are combined. The combined estimate of the transmitted chip sequences is despread to recover transmitted data.

BRIEF DESCRIPTION OF THE DRAWINGS

A more detailed understanding of the invention may be had from the following description of a preferred embodiment, given by way of example and to be understood in conjunction with the accompanying drawings wherein:

FIG. 1 is a block diagram of a transmitter for supporting closed loop mode transmit diversity for dedicated physical channel (DPCH) transmission in accordance with the present invention;

FIGS. 2A and 2B, taken together, are an exemplary block diagram of a receiver implementing B-FFT-based CLE using joint processing with transmit and receive diversity at twice the chip rate in accordance with the present invention;

FIG. 3 shows a space time transmit diversity (STTD) encoder for quadrature phase shift keying (QPSK);

FIG. 4 shows an STTD encoder for 16 quadrature amplitude modulation (16 QAM); and

FIGS. 5A and 5B, taken together, are an exemplary block diagram of a receiver implementing B-FFT-based CLE using joint processing in STTD and receive diversity with over sampling in accordance with the present invention.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

The present invention will be described with reference to the drawing figures wherein like numerals represent like elements throughout.

Hereafter, the terminology “WTRU” includes but is not limited to a user equipment (UE), a mobile station, a fixed or mobile subscriber unit, a pager, or any other type of device capable of operating in a wireless environment.

The features of the present invention may be incorporated into an integrated circuit (IC) or be configured in a circuit comprising a multitude of interconnecting components.

The present invention provides a method and apparatus for implementing an advanced wireless receiver using CLE and joint processing. The joint processing eliminates or reduces mutual channel interference and enhances data detection performance and system throughput. The joint processing-based CLE in accordance with the present invention utilizes transmit diversity and receive diversity with over-sampling. The over-sampling is preferably at twice the chip rate, but the sampling rate may be at any rate. Compared with the receiver using individual equalizers, where each equalizer is dedicated for one antenna, the joint processing-based CLE considers the mutual interference between antennas and eliminates the mutual interferences using joint approaches. Furthermore, the joint processing-based CLE in accordance with the present invention uses B-FFT techniques to realize efficient implementation. The B-FFT and joint processing-based CLE in the present invention has the same number of FFT operations as compared to a prior art receiver without joint processing.

FIG. 1 is a block diagram of a transmitter 100 for supporting closed loop mode transmit diversity for dedicated physical channel (DPCH) transmission in accordance with the present invention. In a closed loop mode transmit diversity, a WTRU sends a feedback signaling message (FSM) to the UMTS terrestrial radio access network (UTRAN) to maximize the received power of the WTRU. Two different closed loop modes, (closed loop modes 1 and 2) are defined. The use of the two closed loop modes is controlled via higher layer signaling.

As shown in FIG. 1, a DPCH data sequence 102, (including a dedicated physical control channel (DPCCH) data sequence and a dedicated physical data channel (DPDCH) data sequence is despread and descrambled by multiplying the DPCH data sequence 102 with a spreading code and scrambling code 104 via a multiplier 106 to generate a spread complex valued signal 108. The spread complex valued signal 108 is fed into multipliers 110, 112, each of which multiplies the spread complex valued signal 108 by a first antenna specific weight factor 114, w₁, and a second antenna specific weight factor 116, w₂, respectively. The weight factors 114, 116 are complex valued signals, (i.e., w_(i)=a_(i)+jb_(i)), which are generated by a weight generator 118 based on a feedback information (FBI) message 120 from an uplink DPCCH.

As shown in FIG. 1, the resulting signals 122, 124 output from the multipliers 110, 112 are respectively summed with respective common pilot channels (CPICHs) 126, 128 via a respective summer 130, 132 to generate transmission signals 134, 136 which are transmitted by respective antennas 138, 140.

The weight factors 114, 116 correspond to phase adjustments in a closed loop mode 1 and phase/amplitude adjustments in a closed loop mode 2. For the closed loop mode 1, different, (preferably orthogonal), dedicated pilot symbols in the DPCCH are transmitted by the antennas 138, 140. For the closed loop mode 2, the same dedicated pilot symbols in the DPCCH are transmitted by the antennas 138, 140.

The transmitter 100 uses the CPICH signals 126, 128 transmitted from the antenna 138 and the antenna 140 to calculate the phase adjustment to be applied at the UTRAN to maximize the received power of a WTRU including the receiver 200 of FIGS. 2A and 2B. In each time slot, the receiver 200 calculates the optimum phase adjustment, φ, for antenna 140, which is then quantized into φ_(Q) having two possible values as follows: $\begin{matrix} {\phi_{Q} = \left\{ {\begin{matrix} {\pi,} & {{{if}\quad{\pi/2}} < {\phi - {\phi_{r}(i)}} \leq {3{\pi/2}}} \\ {0,} & {otherwise} \end{matrix};{where}} \right.} & {{Equation}\quad(1)} \\ {{\phi_{r}(i)} = \left\{ \begin{matrix} {0,} & {{i = 0},2,4,6,8,10,12,14} \\ {{\pi/2},} & {{i = 1},3,5,7,9,11,13} \end{matrix} \right.} & {{Equation}\quad(2)} \end{matrix}$ If φ_(Q)=0, a command ‘0’ is sent to the UTRAN using the FSM_(ph) field and if φ_(Q)=π, a command ‘1’ is sent to the UTRAN using the FSM_(ph) field.

Due to rotation of the constellation at the WTRU in the closed loop mode 1, the UTRAN interprets the received commands according to table 1 which shows the mapping between phase adjustment, φ_(i), and the received feedback command for each uplink slot. TABLE 1 Slot # 0 1 2 3 4 5 6 7 8 9 10 11 12 13 14 FSM 0 0  π/2 0  π/2 0  π/2 0  π/2 0  π/2 0  π/2 0  π/2 0 1 π −π/2 π −π/2 π −π/2 π −π/2 π −π/2 π −π/2 π −π/2 π

The weight 116, w₂, is then calculated by averaging the received phases over 2 consecutive slots as follows: $\begin{matrix} {{w_{2} = {\frac{\sum\limits_{i = {n - 1}}^{n}{\cos\left( \phi_{i} \right)}}{2} + {j\frac{\sum\limits_{i = {n - 1}}^{n}{\sin\left( \phi_{i} \right)}}{2}}}};} & {{Equation}\quad(3)} \end{matrix}$ where φ_(i)ε0,π,π/2,−π/2}. For antenna 1, w₁ is constant w₁=1/√{square root over (2)}.

The phase and amplitude are both adjusted in the closed loop mode 2. The adjustments are based on the commands received in the FSM and are summarized in Tables 2 and 3 for the power and phase adjustments respectively. TABLE 2 FSM_(po) Power_ant1 Power_ant2 0 0.2 0.8 1 0.8 0.2

TABLE 3 FSM_(ph) Phase difference between antennas (radians) 000 π 001  −3π/4    011 −π/2   010 −π/4   110 0 111 π/4 101 π/2 100 3π/4 

Antenna 138 transmits data symbols using weight coefficient w₁ ^((k)) 112 and antenna 140 transmits data symbols using weight coefficient w₂ ^((k)) 116 for the k^(th) channelization code.

The received signal can be expressed as follows: $\begin{matrix} {{\underset{\_}{r} = {{H_{1}\left( {\sum\limits_{k = 1}^{K}{w_{1}^{(k)}{\underset{\_}{s}}_{k}}} \right)} + {H_{2}\left( {\sum\limits_{k = 1}^{K}{w_{2}^{(k)}{\underset{\_}{s}}_{k}}} \right)} + n}};} & {{Equation}\quad(4)} \end{matrix}$ where H₁ and H₂ is the channel response matrix corresponding to the first and second (diversity) transmit antennas, respectively. The transmitted chip sequences are related by the spreading code matrix C as s_(k)=C_(k){right arrow over (d)}_(k) for the k^(th) code. The weighted composite chip sequences are ${\overset{\rightarrow}{t}}_{1} = {{\sum\limits_{k = 1}^{K}{w_{1}^{(k)}{\underset{\_}{s}}_{k}\quad{and}\quad{\overset{\rightarrow}{t}}_{2}}} = {\sum\limits_{k = 1}^{K}{w_{2}^{(k)}{{\underset{\_}{s}}_{k}.}}}}$ Equation (4) can be rewritten as follows: r=H ₁ {right arrow over (t)} ₁ +H ₂ {right arrow over (t)} ₂ +n.   Equation (5)

The weighted composite chip sequences {right arrow over (t)}₁ and {right arrow over (t)}₂ can be demodulated using MMSE solution such that: {circumflex over (t)} =(H ^(H) H+σ ² I)⁻¹ H ^(H) r.   Equation (6) The vector {right arrow over ({circumflex over (t)})} is the estimated composite chip sequences and is expressed by {right arrow over ({circumflex over (t)})}=[{right arrow over (t)}₁,{right arrow over (t)}₂]^(T).

In the presence of receive diversity and over-sampling, the channel response matrix H can be written as follows: $\begin{matrix} {{H = \begin{bmatrix} H_{1,o} \\ H_{1,e} \\ H_{2,o} \\ H_{2,e} \end{bmatrix}};} & {{Equation}\quad(7)} \end{matrix}$ where H_(i,o) and H_(i,e) i=1, . . . N are the channel response matrix of the i^(th) receiving antenna for odd and even sample sequences, respectively. Typically, N=2 for receive diversity and twice the chip rate sampling is used. However, N can be any number and the sampling rate can be any rate. For simplicity and illustration purposes, the present invention will be explained with reference to N=2 and twice the chip rate sampling hereinafter. In the presence of transmit adaptive antennas and receive diversity (N=2) with twice the chip rate over-sampling, the channel response matrix H can be written as follows: $\begin{matrix} {{H = \begin{bmatrix} H_{1,o}^{(1)} & H_{1,o}^{(2)} \\ H_{1,e}^{(1)} & H_{1,e}^{(2)} \\ H_{2,o}^{(1)} & H_{2,o}^{(2)} \\ H_{2,e}^{(1)} & H_{2,e}^{(2)} \end{bmatrix}};} & {{Equation}\quad(8)} \end{matrix}$ where H_(i,o) ^((j)) and H_(i,e) ^((j)) are the channel response matrix of the i^(th) receive antenna and the j^(th) transmit antenna for odd and even sample sequences, respectively.

The estimated data symbols {right arrow over (d)}₁ and {right arrow over (d)}₂ can be simply obtained by multiplying the equalized composite chip sequences with complex conjugate of weights for both antennas, adding them up and despreading the added results as follows: d _(k) =C _(k) ^(H)(w ₁ ^((k)) *{right arrow over (t)} ₁ +w ₂ ^((k))*{right arrow over (t)}₂).   Equation (9)

B-FFT is used to realize the joint processing. H_(i,o) represents the channel response matrix for the i^(th) receive antenna and odd sample sequences and for both transmit antenna. H_(i,o) can be expressed as follows: H _(i,o) =[H _(1,o) ⁽¹⁾ H_(i,o) ⁽²⁾].   Equation (10)

The channel response matrix H_(i,o) can be further expressed in details in terms of channel coefficients as follows: $\begin{matrix} {H_{1,o} = {\begin{bmatrix} h_{1,0} & h_{2,0} & \quad & \quad & \quad & \quad & \quad & \quad & \quad \\ h_{1,1} & h_{2,1} & h_{1,0} & h_{2,0} & \quad & \quad & \quad & \quad & \quad \\ \vdots & \vdots & h_{1,1} & h_{2,1} & \quad & \quad & \quad & \quad & \quad \\ \vdots & \vdots & \vdots & \vdots & \quad & \quad & \quad & \quad & \quad \\ h_{1,{W - 2}} & h_{2,{W - 2}} & \vdots & \vdots & ⋰ & \quad & \quad & \quad & \quad \\ h_{1,{W - 1}} & h_{2,{W - 1}} & h_{1,{W - 2}} & h_{1,{W - 2}} & \quad & ⋰ & \quad & \quad & \quad \\ \quad & \quad & h_{1,{W - 1}} & h_{1,{W - 1}} & \quad & \quad & ⋰ & \quad & \quad \\ \vdots & \vdots & \quad & \quad & \quad & \quad & \quad & \quad & \quad \\ \quad & \vdots & \quad & \quad & \quad & \quad & \quad & h_{1,0} & h_{2,0} \\ \quad & \quad & \quad & \quad & ⋰ & \quad & \quad & h_{1,1} & h_{2,1} \\ \quad & \quad & \quad & \quad & \quad & ⋰ & \quad & \vdots & \vdots \\ \quad & \quad & \quad & \quad & \quad & \quad & ⋰ & \vdots & \vdots \\ \vdots & \vdots & \quad & \quad & \quad & \quad & \quad & h_{1,{W - 2}} & h_{2,{W - 2}} \\ \quad & \quad & \quad & \quad & \quad & \quad & \quad & h_{1,{W - 1}} & h_{2,{W - 1}} \end{bmatrix}.}} & {{Equation}\quad(11)} \end{matrix}$ H_(1,o) is expressed by the channel coefficients with pre-ordering of columns of channel matrix to transform the original matrix into a block circular matrix for channel response matrix H and to enable efficient B-FFT computations. Similarly, H_(2,o), H_(1,e) and H_(2,e) can be expressed in the same form that enables the B-FFT.

Each block is defined as follows: H_(i)=[h_(1,i) h_(2,i)], i==0,1,2, . . . ,W−1. H_(1,o) can then be expressed as follows: $\begin{matrix} {{H_{1,o} = \begin{bmatrix} H_{0} & \quad & \quad & \quad \\ H_{1} & H_{0} & \quad & \quad \\ \vdots & H_{1} & \quad & \quad \\ H_{W - 1} & \vdots & \quad & \quad \\ \quad & H_{W - 1} & \quad & \quad \\ \quad & \quad & ⋰ & \quad \\ \quad & \quad & \quad & H_{0} \\ \quad & \quad & \quad & H_{1} \\ \quad & \quad & \quad & \vdots \\ \quad & \quad & \quad & H_{W - 1} \end{bmatrix}};} & {{Equation}\quad(12)} \end{matrix}$ Where each H_(i) is a matrix of size one by two.

F_((P)) and F_((K)) are B-FFT matrices of size P×P and K×K, respectively. The matrix H_(1,o) can be decomposed by B-FFT in an extended manner as follows: H _(1,o) =F _((P)) ⁻¹ Λ _(H) F _((K))   Equation (13) where F _((P)) =F _(L) I _(P);   Equation (14) and F _((K)) =F _(L)

I_(K); Equation   (15) where F_(L) is the L-point FFT matrix, I_(P) and I_(K) are the identity matrix of size P and K, respectively, and

is a kronecker product. For example, L=256 or 512, P=1 and K=2. It should be noted that the foregoing numbers are provided as an example and any other numbers may be implemented. L is scalable for more efficient implementation. Λ_(H) is a block-diagonal matrix whose diagonal blocks are F_((K))H(:,1: K). Λ_(H)=diag(F _((K)) H(:,1: K)).   Equation (16) Also H_(1,o) ^(H) =F _((K)) ⁻¹Λ*_(H) F _((P));   Equation (17) H _(1,o) ^(H) H _(1,o) =F _((K)) ⁻¹Λ*_(H)Λ_(H) F _((K));   Equation (18) and H _(1,o) ^(H) {right arrow over (r)} _(1,o) =F _((K)) ⁻¹Λ*_(H) F _((P)) {right arrow over (r)} _(1,o).   Equation (19)

The transmitted data sequence s can be solved by the following equations: y=Λ_(H) ^(H) F _((P)) {right arrow over (r)};   Equation (20) y=κ _(H) ^(H)κ_(H) x; and   Equation (21) t=F _((K)) ⁻¹ x.   Equation (22)

In general, x can be solved block by block using Cholesky decomposition. Since the block size is very small (only 2×2) here in consideration, a direct matrix inverse of each block can be performed without using Cholesky decomposition. A similar approach can also be developed using time domain channel correlation matrix R=H^(H)H.

The correlation matrix R can be decomposed by B-FFT as follows: R=F _((P)) ⁻¹Λ_(H) F _((K));   Equation (23) where Λ_(R) is a block-diagonal matrix whose diagonal blocks are F_((K))R(:,1: K)

The above procedure is performed for H_(1,o), H_(2,o), H_(1,e) and H_(2,e) to develop the entire solution of joint processing and B-FFT is used to realize the joint processing for transmit adaptive antenna and receive diversity.

The detected data symbols of two transmit data sequences using joint processing are as follows: $\begin{matrix} {\overset{\rightarrow}{t} = {\left\lbrack {{\sum\limits_{i = 1}^{N}\left( {{H_{i,o}^{H}H_{i,o}} + {H_{i,e}^{H}H_{i,e}}} \right)} + {\sigma^{2}I}} \right\rbrack^{- 1} \cdot {\left\lbrack {{\sum\limits_{i = 1}^{N}{H_{i,o}^{H}{\overset{\rightarrow}{r}}_{i,o}}} + {H_{i,e}^{H}{\overset{\rightarrow}{r}}_{i,e}}} \right\rbrack.}}} & {{Equation}\quad(24)} \end{matrix}$

The realization of joint processing using B-FFT are as follows: $\begin{matrix} {\overset{\rightarrow}{t} = {{F_{(K)}^{- 1}\left\lbrack {{\sum\limits_{i = 1}^{N}\left( {{\Lambda_{i,o}^{*}\Lambda_{i,o}} + {\Lambda_{i,e}^{*}\Lambda_{i,e}}} \right)} + {\sigma^{2}I}} \right\rbrack}^{- 1} \cdot {\quad{\left\lbrack {{\sum\limits_{i = 1}^{N}{\Lambda_{i,o}^{*}F_{(P)}{\overset{\rightarrow}{r}}_{i,o}}} + {\Lambda_{i,e}^{*}F_{(P)}{\overset{\rightarrow}{r}}_{i,e}}} \right\rbrack.}}}} & {{Equation}\quad(25)} \end{matrix}$

By letting T and {right arrow over (y)} represent as follows: $\begin{matrix} {{{T = {{\sum\limits_{i = 1}^{N}\left( {{\Lambda_{i,o}^{*}\Lambda_{i,o}} + {\Lambda_{i,e}^{*}\Lambda_{i,e}}} \right)} + {\sigma^{2}I}}};}{{and},}} & {{Equation}\quad(26)} \\ {{\overset{\rightarrow}{y} = {{\sum\limits_{i = 1}^{N}{\Lambda_{i,o}^{*}F_{(P)}{\overset{\rightarrow}{r}}_{i,o}}} + {\Lambda_{i,e}^{*}F_{(P)}{\overset{\rightarrow}{r}}_{i,e}}}};} & {{Equation}\quad(27)} \end{matrix}$ Equation (25) can be rewritten as follows: T·F _((K)) {right arrow over (t)}={right arrow over (y)}.   Equation (28)

{right arrow over (x)}=F_((K)){right arrow over (t)} by Equation (22). Therefore, Equation (27) can be rewritten as follows: T{right arrow over (x)}={right arrow over (y)}.   Equation (29)

The unknown {right arrow over (x)} is solved first. Once {right arrow over (x)} are solved, inverse FFT is performed on {right arrow over (x)} to obtain the composite chip sequences to be estimated as follows: {right arrow over (t)}=F _((K)) ⁻¹ {right arrow over (x)}.   Equation (30)

F_((K)) ⁻¹ is exchangeable with F_((K)) as follows: $\begin{matrix} {F_{(k)}^{- 1} = {\frac{1}{L}{F_{(K)}^{*}.}}} & {{Equation}\quad(31)} \end{matrix}$

FIGS. 2A and 2B, taken together, are an exemplary block diagram of a receiver 200 implementing B-FFT-based CLE using joint processing with transmit and receive diversity with two transmit antennas and two receive antennas at twice the chip rate in accordance with the present invention. As explained hereinbefore, any number of transmit and receive antennas and any sampling rate may be used. In this example, for a received signal r, four sample streams 202 ₁-202 ₄ are generated from two receive antennas (not shown). From the sample streams 202 ₁-202 ₄, channel responses between a first transmit antenna and two receive antennas for even and odd sample sequences {right arrow over (h)}⁽¹⁾ 206 ₁-206 ₄ and channel responses between a second transmit antenna and two receive antennas for even and odd sample sequences {right arrow over (h)}⁽²⁾ 206 ₅-206 ₈ are generated by a channel estimator (not shown).

The sample streams 202 ₁-202 ₄ are processed by FFT units 204 ₁-204 ₄ to be converted into frequency domain data, respectively. The channel response vectors 206 ₁-206 ₈ are processed by FFT units 208 ₁-208 ₈, respectively to generate frequency domain channel response vectors 210 ₁-210 ₈. Complex conjugates 214 ₁-214 ₈ of the frequency domain channel response vectors 210 ₁-210 ₈ are generated by complex conjugate units 212 ₁-212 ₈, respectively. The frequency domain sample streams 216 ₁-216 ₄ and complex conjugates 214 ₁-214 ₈ of the frequency domain channel response vectors 210 ₁-210 ₈ are multiplied by element-wise multipliers 218 ₁-218 ₈, respectively. The multiplication results for the first transmit antenna 220 ₁-220 ₄ are combined by a combiner 222 ₁ and the multiplication results for the second transmit antenna 220 ₅-220 ₈ are combined by a combiner 222 ₂. The combined results y⁽¹⁾, y⁽²⁾, (224 ₁, 224 ₂), correspond to the output of Equation (20) (or Equation (27)).

The frequency domain channel response vectors 210 ₁-210 ₈ and a noise variance value 232 enter a joint channel correlation generator 230. Equation (18) depicts the function of generator 230 for channel correlation generation that occurs in frequency domain. The function of processor 240 is depicted by Equations (18), (20), (21) and (22) for solving the linear systems. The joint channel correlation generator 230 generates joint channel correlation matrix 234 ₁-234 ₄ between two transmit antennas and two receive antennas and even and odd sample stream. The joint channel correlation matrixes 234 ₁-234 ₄ are combined by a combiner 236 and the combined joint channel correlation matrix 238, which corresponds to T in Equation (26), enters a processor 240.

The processor 240 receives as an input the combined joint channel correlation matrix 238 and two combined results y⁽¹⁾, y⁽²⁾ 224 ₁, 224 ₂ and generates estimates of the transmitted chip sequences by solving the 2×2 linear systems of Equation (29). The estimates of the transmitted chip sequences 242 ₁, 242 ₂ undergo transmit adaptive antenna processing by being multiplied with complex conjugates 248 ₁, 248 ₂ of weight factors 244 ₁, 244 ₂ generated by complex conjugate units 246 ₁, 246 ₂, by element-wise multipliers 218 ₉ and 218 ₁₀, respectively. The two multiplier outputs 250 ₁, 250 ₂ are soft combined by a summer 252 and the combined output 254 is processed by an IFFT unit 256 to be converted into time domain signals 258. Then, the time domain signals 258 are processed by a despreader 260 to generate a data symbol estimate 262.

The present invention may be implemented with STTD. For STTD, a first antenna transmits {right arrow over (d)}₁ and a second antenna transmits {right arrow over (d)}₂, where {right arrow over (d)}₁ and {right arrow over (d)}₂ are STTD encoded data sequences. FIG. 3 shows the STTD encoded data sequences for QPSK, such that {right arrow over (d)}₁=[b₀ b₁ b₂ b₃]^(T) and {right arrow over (d)}₂=[{overscore (b)}₂ b₃ b₀ {overscore (b)}₁]^(T). FIG. 4 shows the STTD encoded data sequences for 16 QAM such that {right arrow over (d)}₁=[b₀ b₁ b₂ b₃ b₄ b₅ b₆ b₇]^(T) and {right arrow over (d)}₂=[{overscore (b)}₄ b₅ b₆ b₇ b₀{overscore (b)}₁ b₂ b₃]^(T).

The received signal at the receiver can be expressed as follows: r=H ₁ s ₁ +H ₂ s ₂ +n;   Equation (32) where H₁ and H₂ is the channel response matrix corresponding to the first and second diversity antennas, respectively. The chip and STTD encoded symbol sequences are related by the spreading code matrix C as s₁=C{right arrow over (d)}₁ and s₂=C{right arrow over (d)}₂.

The chip sequences s₁ and s₂ can be demodulated at the receiver using MMSE such that: ŝ=(H ^(H) H+σ ² I)⁻¹ H ^(H) r.   Equation (33)

In the presence of receive diversity and over-sampling, the channel response matrix H can be expressed by Equation (7) and in the presence of STTD transmit diversity and receive diversity (N=2) with twice the chip rate over-sampling, the channel response matrix H can be expressed by Equation (8).

The STTD encoded data symbols {right arrow over (d)}₁ and {right arrow over (d)}₂ can be simply obtained by de-spreading the equalized chip sequences. Because data sequences b_(i), i=0,1,2, . . . ,7 are detected in both STTD encoded data vectors {right arrow over (d)}₁ and {right arrow over (d)}₂, the STTD decoding and soft combining are used to achieve diversity gain and improve performance such as: d=α ₁·sign(b _(i,ant1))+α₂·sign(b _(i,ant2));   Equation (34) where the notation sign ( ) represents the sign changes according to STTD decoding rules and modulation types, such as QPSK and 16 QAM.

For QPSK, the STTD decoding are described as follows:

Antenna 1: sign(b _(i,ant1))=b _(i,ant1), for all i

Antenna 2: sign(b _(i,ant2))=b _(i,ant2), if i=0,3 sign(b _(i,ant2))=−b _(i,ant2), else (or i=1,2)

For 16 QAM, the STTD decoding are as follows:

Antenna 1: sign(b _(i,ant1))=b _(i,ant1), for all i

Antenna 2: sign(b _(i,ant1))=b _(i,ant2), if i=0,2,3,5,6,7 sign(b _(i,ant2))=−b _(i,ant2), else (or i=1,4)

For equal gain soft combining, the weight coefficients are α=α₂=1. For maximal ratio combining (MRC), the weight coefficients α_(n), n=1,2 are preferably as follows: $\begin{matrix} {{\alpha_{n} = \sqrt{\frac{\sum\limits_{i}{h_{n,i}}^{2}}{{\sum\limits_{i}{h_{1,i}}^{2}} + {\sum\limits_{i}{h_{2,i}}}}}},{n = 1},2.} & {{Equation}\quad(35)} \end{matrix}$

B-FFT is used to realize the joint processing. For example, H_(i,o) representing the channel response matrix for the i^(th) receive antenna and odd sampled sequences and for both transmit antenna can be expressed as follows: H _(i,o) =[H _(i,o) ⁽¹⁾ H_(i,o) ⁽²⁾].   Equation (36)

The channel response matrix H_(i,o) can be expressed in details by Equation (11) in terms of channel coefficients and can also be expressed by Equation (12). The matrix H_(i,o) can be decomposed by B-FFT by Equations (13)-(15).

The transmitted data sequence s can be estimated by the following equations: y=F _((P)) r;   Equation (37) Λ_(H) ^(H) y=Λ _(H) ^(H)Λ_(H) x;   Equation (38) s=F _((K)) ⁻¹ x.   Equation (39)

In general, the x can be solved block by block using Cholesky decomposition. Since the block size is very small (only 2×2) for the example under consideration, a solution using a direct matrix inverse of each block can be performed without using Cholesky decomposition. A similar approach can also be developed using a time domain channel correlation matrix R=H^(H)H. Same procedure is repeated for H_(1,o), H_(2,o), H_(1,e) and H_(2,e) to develop the entire solution of joint processing and B-FFT is used to realize the joint processing for STTD and receive diversity.

The detected data symbols of two transmit data sequences using joint processing are expressed as follows: $\begin{matrix} {{\overset{->}{d}}_{Tx} = {\left\lbrack {{\sum\limits_{i = 1}^{N}\left( {{H_{i,o}^{H}H_{i,o}} + {H_{i,e}^{H}H_{i,e}}} \right)} + {\sigma^{2}I}} \right\rbrack^{- 1} \cdot {\quad{\left\lbrack {{\sum\limits_{i = 1}^{N}{H_{i,o}^{H}{\overset{->}{r}}_{i,o}}} + {H_{i,e}^{H}{\overset{->}{r}}_{i,e}}} \right\rbrack.}}}} & {{Equation}\quad(40)} \end{matrix}$

The realization of joint processing using B-FFT are as follows: $\begin{matrix} {{\overset{->}{d}}_{Tx} = {{F_{(K)}^{- 1}\left\lbrack {{\sum\limits_{i = 1}^{N}\left( {{\Lambda_{i,o}^{*}\Lambda_{i,o}} + {\Lambda_{i,e}^{*}\Lambda_{i,e}}} \right)} + {\sigma^{2}I}} \right\rbrack}^{- 1} \cdot {\quad{\left\lbrack {{\sum\limits_{i = 1}^{N}{\Lambda_{i,o}^{*}F_{(P)}{\overset{\rightarrow}{r}}_{i,o}}} + {\Lambda_{i,e}^{*}F_{(P)}{\overset{\rightarrow}{r}}_{i,e}}} \right\rbrack.}}}} & {{Equation}\quad(41)} \end{matrix}$

By letting R_(fft) and {right arrow over (y)} represent as follows: $\begin{matrix} {{{R_{fft} = {{\sum\limits_{i = 1}^{N}\left( {{\Lambda_{i,o}^{*}\Lambda_{i,o}} + {\Lambda_{i,e}^{*}\Lambda_{i,e}}} \right)} + {\sigma^{2}I}}};}{and}} & {{Equation}\quad(42)} \\ {{\overset{\rightarrow}{y} = {{\sum\limits_{i = 1}^{N}{\Lambda_{i,o}^{*}F_{(P)}{\overset{\rightarrow}{r}}_{i,o}}} + {\Lambda_{i,e}^{*}F_{(P)}{\overset{\rightarrow}{r}}_{i,e}}}},} & {{Equation}\quad(43)} \end{matrix}$ the Equation (41) can be rewritten as follows: R _(fft) F _((K)) {right arrow over (d)} _(Tx) ={right arrow over (y)}.   Equation (44)

Furthermore, by letting {right arrow over (x)}=F_((K)){right arrow over (d)}_(Tx), a linear system is obtained such that: R_(fft) {right arrow over (x)}={right arrow over (y)}.   Equation (45)

After solving the unknown {right arrow over (x)}, an inverse FFT is performed on {right arrow over (x)} to obtained the data symbols to be estimated as follows: {right arrow over (d)} _(Tx) =F _((K)) ⁻¹ {right arrow over (x)}.   Equation (46)

F_((K)) ⁻¹ is exchangeable with F_((K)) as follows: $\begin{matrix} {F_{(K)}^{- 1} = {\frac{1}{L}{F_{(K)}^{*}.}}} & {{Equation}\quad(47)} \end{matrix}$

FIGS. 5A and 5B, taken together, are a block diagram of a receiver 300 implementing B-FFT-based CLE using joint processing in STTD and receive diversity with over sampling in accordance with the present invention. As explained hereinbefore, any number of transmit and receive antennas and any sampling rate may be used. In this example, for a received signal r, four sample streams 302 ₁-302 ₄ are generated from two receive antennas (not shown). From the sample streams 302 ₁-302 ₄, channel responses between a first transmit antenna and two receive antennas for even and odd sample sequences {right arrow over (h)}⁽¹⁾ 306 ₁-306 ₄ and channel responses between a second transmit antenna and two receive antennas for even and odd sample sequences {right arrow over (h)}⁽²⁾ 306 ₅-306 ₈ are generated by a channel estimator (not shown).

The sample streams 302 ₁-302 ₄ are processed by FFT units 304 ₁-304 ₄ to be converted into frequency domain data, respectively. The channel response vectors 306 ₁-306 ₈ are processed by FFT units 308 ₁-308 ₈, respectively to generate frequency domain channel response vectors 310 ₁-310 ₈. Complex conjugates 314 ₁-314 ₈ of the frequency domain channel response vectors 310 ₁-310 ₈ are generated by complex conjugate units 312 ₁-312 ₈, respectively. The frequency domain sample streams 316 ₁-316 ₄ and complex conjugates 314 ₁-314 ₈ of the frequency domain channel response vectors 310 ₁-310 ₈ are multiplied by element-wise multipliers 318 ₁-318 ₈, respectively. The multiplication results for the first transmit antenna 320 ₁-320 ₄ are combined by a combiner 322 ₁ and the multiplication results for the second transmit antenna 320 ₅-320 ₈ are combined by a combiner 322 ₂. The combined results y⁽¹⁾, y⁽²⁾, (324 ₁, 324 ₂), which correspond to the output of Equation (48).

The frequency domain channel response vectors 310 ₁-310 ₈ and a noise variance value 332 enter a joint channel correlation generator 330. Equation (18) depicts the function of generator 330. Equations (38), (39) and (40) depict the function of processor 340. The joint channel correlation generator 330 generates joint channel correlation matrix 334 ₁-334 ₄ between two transmit antennas and two receive antennas for even and odd sample streams. The joint channel correlation matrixes 334 ₁-334 ₄ are combined by a combiner 336 and the combined joint channel correlation matrix 338, which corresponds to R_(fft) in Equation (42), enters a processor 340.

The processor 340 receives as an input the combined joint channel correlation matrix 338 and two combined results y⁽¹⁾, y⁽²⁾, 324 ₁, 324 ₂ and generates estimates of the transmitted chip sequences by solving the 2×2 linear systems of Equation (45). The equalized chip sequences 342 ₁, 342 ₂ are STTD decoded and soft combined by a STTD decoder/soft combiner 350 as shown in Equation (34). The STTD decoded and combined chip sequences 352 is processed by an IFFT unit 354 and despreader 356 to generate an estimate of transmitted data 358.

Although the features and elements of the present invention are described in the preferred embodiments in particular combinations, each feature or element can be used alone without the other features and elements of the preferred embodiments or in various combinations with or without other features and elements of the present invention. 

1. In a wireless communication system including a transmitter having a plurality of antennas for transmission and a receiver having a plurality of antennas for reception, a method of performing chip level equalization (CLE) using joint processing of received signals, the method comprising: generating a sample sequence from received signals; generating a channel response matrix between the plurality of transmit antennas and the plurality of receive antennas from the sample sequence; generating a joint channel correlation matrix between the transmit antennas and the receive antennas using a block fast Fourier transform (B-FFT) decomposition of the channel response matrix; generating estimates of transmitted chip sequences from each transmit antenna using minimum mean square error (MMSE) and the joint channel correlation matrix; combining the estimates of transmitted chip sequences from the transmit antennas; and despreading the combined estimate of transmitted chip sequences.
 2. The method of claim 1 wherein a closed loop mode transmit diversity is implemented.
 3. The method of claim 2 further comprising: multiplying complex conjugate of a weight to the estimates of transmitted chip sequences, the weight being applied to a transmitted chip sequence at a transmitter for the closed loop mode transmit diversity.
 4. The method of claim 3 wherein the closed loop mode transmit diversity is either mode 1 or mode
 2. 5. The method of claim 1 wherein a space time transmit diversity (STTD) is implemented in transmission.
 6. The method of claim 5 further comprising a step of an STTD decoding of the estimates of transmitted chip sequences.
 7. The method of claim 1 wherein the estimates of the transmitted chip sequence is performed by using Cholesky decomposition block by block.
 8. The method of claim 1 wherein the estimates of the transmitted chip sequence is performed by direct matrix inversion.
 9. The method of claim 1 wherein the received signals are over-sampled.
 10. The method of claim 9 wherein the received signals are over-sampled at twice the chip rate.
 11. In a wireless communication system including a transmitter having a plurality of antennas for transmission and a receiver having a plurality of antennas for reception, an apparatus for performing chip level equalization (CLE) using joint processing of received signals, the apparatus comprising: a sampling unit for generating a sample sequence from received signals; a channel estimator for generating a channel response matrix between the plurality of transmit antennas and the plurality of receive antennas from the sample sequence; a joint channel correlation generator for generating a joint channel correlation matrix between the transmit antennas and the receive antennas using a block fast Fourier transform (B-FFT); a processing unit for generating estimates of transmitted chip sequences from each transmit antenna using minimum mean square error (MMSE) and a B-FFT based on the joint channel correlation matrix; a soft combiner for combining the estimates of transmitted chip sequences from each transmit antenna; an inverse fast Fourier transform (IFFT) unit for performing IFFT on a combined estimates from the combiner; and a despreader for despreading an output of the IFFT unit.
 12. The apparatus of claim 11 wherein the processing unit comprising: a plurality of fast Fourier transform (FFT) units for performing FFT on the samples; a plurality of FFT units for performing FFT on channel impulse responses between each transmit antenna and each receive antenna; a plurality of complex conjugate generators for generating complex conjugate of the FFT on channel impulse responses; a plurality of multipliers for multiplying the FFT on the samples and the complex conjugate of the FFT on channel impulse responses; a plurality of combiners for combining the multiplication results corresponding each of the transmit antennas; and a processor for generating estimates of transmitted chip sequences from each transmit antenna from the results of the combiners and the joint channel correlation matrix.
 13. The apparatus of claim 11 wherein closed loop mode transmit diversity is implemented.
 14. The apparatus of claim 13 further comprising: a plurality of complex conjugate generators for generating complex conjugates of weights applied to the transmit antennas for the closed loop mode transmit diversity; and a plurality of multipliers for multiplying a complex conjugate of a weight to the estimates of transmitted chip sequence corresponding each transmit antenna, respectively.
 15. The apparatus of claim 13 wherein the closed loop mode transmit diversity is either mode 1 or mode
 2. 16. The apparatus of claim 11 wherein a space time transmit diversity (STTD) is implemented in transmission.
 17. The apparatus of claim 16 further comprising an STTD decoder for performing STTD decoding of the estimates of transmitted chip sequences.
 18. The apparatus of claim 11 wherein the estimates of the transmitted chip sequence is performed by using Cholesky decomposition block by block.
 19. The apparatus of claim 11 wherein the estimates of the transmitted chip sequence is performed by direct matrix inversion.
 20. The apparatus of claim 11 wherein the received signals are over-sampled.
 21. The apparatus of claim 20 wherein the received signals are over-sampled at twice the chip rate.
 22. In a wireless communication system including a transmitter having a plurality of antennas for transmission and a receiver having a plurality of antennas for reception, an integrated circuit (IC) for performing chip level equalization (CLE) using joint processing of received signals, the IC comprising: a sampling unit for generating a sample sequence from received signals; a channel estimator for generating a channel response matrix between the plurality of transmit antennas and the plurality of receive antennas from the sample sequence; a joint channel correlation generator for generating a joint channel correlation matrix between the transmit antennas and the receive antennas using a block fast Fourier transform (B-FFT); a processing unit for generating estimates of transmitted chip sequences from each transmit antenna using minimum mean square error (MMSE) and a B-FFT based on the joint channel correlation matrix; a soft combiner for combining the estimates of transmitted chip sequences from each transmit antenna; an inverse fast Fourier transform (IFFT) unit for performing IFFT on a combined estimates from the combiner; and a despreader for despreading an output of the IFFT unit.
 23. The IC of claim 22 wherein the processing unit comprising: a plurality of fast Fourier transform (FFT) units for performing FFT on the samples; a plurality of FFT units for performing FFT on channel impulse responses between each transmit antenna and each receive antenna; a plurality of complex conjugate generators for generating complex conjugate of the FFT on channel impulse responses; a plurality of multipliers for multiplying the FFT on the samples and the complex conjugate of the FFT on channel impulse responses; a plurality of combiners for combining the multiplication results corresponding each of the transmit antennas; and a processor for generating estimates of transmitted chip sequences from each transmit antenna from the results of the combiners and the joint channel correlation matrix.
 24. The IC of claim 22 wherein closed loop mode transmit diversity is implemented.
 25. The IC of claim 24 further comprising: a plurality of complex conjugate generators for generating complex conjugates of weights applied to the transmit antennas for the closed loop mode transmit diversity; and a plurality of multipliers for multiplying a complex conjugate of a weight to the estimates of transmitted chip sequence corresponding each transmit antenna, respectively.
 26. The IC of claim 24 wherein the closed loop mode transmit diversity is either mode 1 or mode
 2. 27. The IC of claim 22 wherein a space time transmit diversity (STTD) is implemented in transmission.
 28. The IC of claim 27 further comprising an STTD decoder for performing STTD decoding of the estimates of transmitted chip sequences.
 29. The IC of claim 22 wherein the estimates of the transmitted chip sequence is performed by using Cholesky decomposition block by block.
 30. The IC of claim 22 wherein the estimates of the transmitted chip sequence is performed by direct matrix inversion.
 31. The IC of claim 22 wherein the received signals are over-sampled.
 32. The IC of claim 31 wherein the received signals are over-sampled at twice the chip rate. 